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Devices with very high circuit efficiency. Which Stirling engine has the best design with maximum efficiency. Examples of successful implementation of automobile Stirlings

This article will talk about the familiar, but many do not understand, term coefficient useful action(efficiency). What is it? Let's figure it out. Efficiency factor, hereinafter referred to as efficiency, is a characteristic of the efficiency of the system of any device in relation to the conversion or transmission of energy. It is determined by the ratio of the useful energy used to the total amount of energy received by the system. Is it usually indicated? (" this"). ? = Wpol/Wcym. Efficiency is a dimensionless quantity and is often measured as a percentage. Mathematically determination of efficiency can be written in the form: n=(A:Q) x100%, where A is useful work, and Q is expended work. Due to the law of conservation of energy, efficiency is always less than one or equal to it, that is, it is impossible to obtain useful work more than the energy expended! Looking through different sites, I am often surprised how radio amateurs report, or rather, praise their designs for high efficiency, without having any idea what it is! For clarity, let’s look at a simplified converter circuit using an example and find out how to find the efficiency of the device. A simplified diagram is shown in Fig. 1

Let's say we took as a basis a step-up DC/DC voltage converter (hereinafter referred to as PN), from unipolar to increased unipolar. We connect the ammeter RA1 into the power supply circuit break, and the voltmeter RA2 parallel to the power supply input PN, the readings of which are needed to calculate the power consumption (P1) of the device and the load together from the power source. At the output of the PN in the load supply break we also connect an ammeter RAZ and a voltmeter RA4, which are required to calculate the power consumed by the load (P2) from the PN. So, everything is ready to calculate the efficiency, then let's get started. We turn on our device, take measurements of instrument readings and calculate the powers P1 and P2. Hence P1=I1 x U1, and P2=I2 x U2. Now we calculate the efficiency using the formula: efficiency (%) = P2: P1 x100. Now you have found out approximately the real efficiency of your device. Using a similar formula, you can calculate PN with a two-polar output using the formula: Efficiency (%) = (P2+P3) : P1 x100, as well as a step-down converter. It should be noted that the value (P1) also includes the current consumption, for example: a PWM controller, and (or) a driver for controlling field-effect transistors, and other design elements.

For reference: car amplifier manufacturers often indicate the output power of the amplifier is much higher than in reality! But you can find out the approximate real power of a car amplifier using a simple formula. Let’s say there is a +12v fuse on the car amplifier in the power supply circuit, there is a 50 A fuse. We calculate, P = 12V x 50A, and in total we get a power consumption of 600 W. Even in high-quality and expensive models The efficiency of the entire device is unlikely to exceed 95%. After all, part of the efficiency is dissipated in the form of heat on powerful transistors, transformer windings, and rectifiers. So let's go back to the calculation, we get 600 W: 100% x92=570W. Consequently, this car amplifier will not produce any 1000 W or even 800 W, as the manufacturers write! I hope this article will help you understand such a relative value as efficiency! Good luck to everyone in developing and repeating designs. The invertor was with you.

The beetle is assembled according to the Hartley scheme with a non-standard inclusion feedback, due to which the efficiency is 10-20% higher than similar circuits. This circuit is similar to that used in the simplest telephone bug. It has been circulating on the Internet for a long time, and site owners continue to copy it from each other, not noticing the gross error in the scheme. This error has been fixed here.

R1=R3=R4 - 9.1 k,
R2 - 300 k,
C1 - 0.1 µF,
S2 - 56, S3 - 24,
VT1 - KT315,
VT2 - KT325VM,
L1 - 5+5 turns
wires PEV-0.5
on a 3mm mandrel.

As a rule, the circuit starts working immediately after assembly. If a squeak is heard in the receiver, you should bypass the circuit with a capacitor with a capacity of at least 1 µF. It is better to connect the antenna through a conductor with a capacity of 1-2 pf. With an antenna length of 20cm, my range was 140m.

Photos of the finished device in the version powered by 2 lithium tablets CR-1220 (6v). (work for a very long time):

List of radioelements

Designation	Type	Denomination	Quantity	Note	My notepad
VT1	Bipolar transistor	KT315A	1		To notepad
VT2	Transistor	KT325VM	1		To notepad
C1	Capacitor	0.1 µF	1		To notepad
C2	Capacitor	56 pF	1		To notepad
C3	Capacitor	24 pF	1		To notepad
	Capacitor	1-2 pF	1	To connect the antenna	To notepad
	Capacitor	Not less than 1 µF	1	To bypass the circuit	To notepad
R1, R3, R4	Resistor	9.1 kOhm	3		To notepad
R2	Resistor	300 kOhm	1		To notepad
L1	Inductor		1

The described device provides exceptionally high conversion efficiency, allows regulation of the output voltage and its stabilization, and operates stably when the load power varies. This type of converter is interesting and undeservedly little widespread - quasi-resonant, which is largely free from the disadvantages of other popular circuits. The idea of creating such a converter is not new, but practical implementation became feasible relatively recently, after the advent of powerful high-voltage transistors that allow significant pulse collector current at a saturation voltage of about 1.5 V. Home distinctive feature and the main advantage of this type of power source is high Converter efficiency voltage, reaching 97...98% without taking into account losses in the secondary circuit rectifier, which are mainly determined by the load current.

The quasi-resonant converter differs from a conventional pulse converter, in which by the moment the switching transistors are closed, the current flowing through them is maximum, the quasi-resonant one differs in that by the moment the transistors are closed, their collector current is close to zero. Moreover, the reduction in current at the moment of closing is ensured by the reactive elements of the device. It differs from resonant in that the conversion frequency is not determined by the resonant frequency of the collector load. Thanks to this, it is possible to regulate the output voltage by changing the conversion frequency and realize stabilization of this voltage. Since by the time the transistor closes, the reactive elements reduce the collector current to a minimum, the base current will also be minimal and, therefore, the closing time of the transistor is reduced to the value of its opening time. Thus, the problem of through current that occurs during switching is completely eliminated. In Fig. 4.22 shown circuit diagram self-generating unstabilized power supply.

Main technical characteristics:

Overall efficiency of the unit, %................................................... ....................92;

Output voltage, V, with a load resistance of 8 Ohms....... 18;

Operating frequency of the converter, kHz....................................20;

Maximum output power, W.........................................55;

Maximum amplitude of output voltage ripple with operating frequency, V

The main share of power losses in the unit falls on the heating of the rectifier diodes of the secondary circuit, and the efficiency of the converter itself is such that there is no need for heat sinks for transistors. The power losses on each of them does not exceed 0.4 W. Special selection Transistors are also not required for any parameters. When the output is shorted or the maximum output power is exceeded, generation is interrupted, protecting the transistors from overheating and breakdown.

The filter, consisting of capacitors C1...SZ and inductor LI, L2, is designed to protect the supply network from high-frequency interference from the converter. The autogenerator is started by circuit R4, C6 and capacitor C5. The generation of oscillations occurs as a result of the action of positive feedback through transformer T1, and their frequency is determined by the inductance of the primary winding of this transformer and the resistance of resistor R3 (as the resistance increases, the frequency increases).

Chokes LI, L2 and transformer T1 are wound on identical ring magnetic cores K12x8x3 made of 2000NM ferrite. The inductor windings are performed simultaneously, “in two wires,” using PELSHO-0.25 wire; number of turns - 20. Winding I of the TI transformer contains 200 turns of PEV-2-0.1 wire, wound in bulk, evenly around the entire ring. Windings II and III are wound “in two wires” - 4 turns of PELSHO-0.25 wire; winding IV is a turn of the same wire. For the T2 transformer, a K28x16x9 ring magnetic core made of 3000NN ferrite was used. Winding I contains 130 turns of PELI10-0.25 wire, laid turn to turn. Windings II and III - 25 turns of PELSHO-0.56 wire each; winding - “in two wires”, evenly around the ring.

Choke L3 contains 20 turns of PELI10-0.25 wire, wound on two folded together ring magnetic cores K12x8x3 made of 2000NM ferrite. Diodes VD7, VD8 must be installed on heat sinks with a dissipation area of at least 2 cm2 each.

The described device was developed for use in conjunction with analog stabilizers on different meanings voltage, so there was no need for deep suppression of ripple at the output of the unit. Ripple can be reduced to the required level by using LC filters that are common in such cases, such as, for example, in another version of this converter with the following basic technical characteristics:

Rated output voltage, V................................................... 5,

Maximum output current, A................................................... ......... 2;

Maximum pulsation amplitude, mV............................................50;

Change in output voltage, mV, no more, when the load current changes

from 0.5 to 2 A and mains voltage from 190 to 250 V........................150;

Maximum conversion frequency, kHz.................................... 20.

The circuit of a stabilized power supply based on a quasi-resonant converter is shown in Fig. 4.23.

The output voltage is stabilized by a corresponding change in the operating frequency of the converter. As in the previous block, powerful transistors VT1 and VT2 do not need heat sinks. Symmetrical control of these transistors is implemented using a separate master pulse generator assembled on a DDI chip. Trigger DD1.1 operates in the generator itself.

The pulses have a constant duration specified by the circuit R7, C12. The period is changed by the OS circuit, which includes optocoupler U1, so that the voltage at the output of the unit is maintained constant. The minimum period is set by circuit R8, C13. The DDI.2 trigger divides the repetition frequency of these pulses by two, and the square wave voltage is supplied from the direct output to transistor amplifier current VT4, VT5. Next, the current-amplified control pulses are differentiated by the circuit R2, C7, and then, already shortened to a duration of approximately 1 μs, they enter through the transformer T1 into the base circuit of transistors VT1, VT2 of the converter. These short pulses They serve only to switch transistors - closing one of them and opening the other.

In addition, the main power from the excitation generator is consumed only when switching powerful transistors, so the average current consumed by it is small and does not exceed 3 mA, taking into account the current of the zener diode VD5. This allows it to be powered directly from the primary network through the quenching resistor R1. Transistor VT3 is a control signal voltage amplifier, as in a compensation stabilizer. The stabilization coefficient of the block's output voltage is directly proportional to the static current transfer coefficient of this transistor.

The use of transistor optocoupler U1 ensures reliable galvanic isolation of the secondary circuit from the network and high noise immunity at the control input of the master oscillator. After the next switching of transistors VT1, VT2, the capacitor SY begins to recharge and the voltage at the base of the transistor VT3 begins to increase, the collector current also increases. As a result, the optocoupler transistor opens, maintaining the master oscillator capacitor C13 in a discharged state. After the rectifier diodes VD8, VD9 are closed, the capacitor SY begins to discharge to the load and the voltage across it drops. Transistor VT3 closes, as a result of which capacitor C13 begins charging through resistor R8. As soon as the capacitor is charged to the switching voltage of the trigger DD1.1, its direct output will be set to high level voltage. At this moment, the next switching of transistors VT1, VT2 occurs, as well as the discharge of the SI capacitor through the opened optocoupler transistor.

The next process of recharging the capacitor SY begins, and the trigger DD1.1 after 3...4 μs will return to the zero state again due to the small time constant of the circuit R7, C12, after which the entire control cycle is repeated, regardless of which of the transistors is VT1 or VT2 - open during the current half-term. When the source is turned on, at the initial moment, when the capacitor SY is completely discharged, there is no current through the optocoupler LED, the generation frequency is maximum and is determined mainly by the time constant of the circuit R8, C13 (the time constant of the circuit R7, C12 is several times smaller). With the ratings of these elements indicated in the diagram, this frequency will be about 40 kHz, and after it is divided by the DDI.2 trigger - 20 kHz. After charging the capacitor SY to the operating voltage, the OS stabilizing loop on the elements VD10, VT3, U1 comes into operation, after which the conversion frequency will already depend on the input voltage and load current. Voltage fluctuations on the capacitor SY are smoothed out by filter L4, C9. Chokes LI, L2 and L3 are the same as in the previous block.

Transformer T1 is made on two ring magnetic cores K12x8x3 folded together from 2000NM ferrite. The primary winding is wound in bulk evenly throughout the entire ring and contains 320 turns of PEV-2-0.08 wire. Windings II and III each contain 40 turns of wire PEL1110-0.15; they are wound “in two wires”. Winding IV consists of 8 turns of PELSHO-0.25 wire. Transformer T2 is made on a ring magnetic core K28x16x9 made of 3000NN ferrite. Winding I - 120 turns of PELSHO-0.15 wire, and II and III - 6 turns of PEL1110-0.56 wire, wound “in two wires”. Instead of PELSHO wire, you can use PEV-2 wire of the appropriate diameter, but in this case it is necessary to lay two or three layers of varnished cloth between the windings.

Choke L4 contains 25 turns of wire PEV-2-0.56, wound on a ring magnetic core K12x6x4.5 made of 100NNH1 ferrite. Any ready-made inductor with an inductance of 30...60 μH for a saturation current of at least 3 A and an operating frequency of 20 kHz is also suitable. All fixed resistors are MJIT. Resistor R4 - adjusted, of any type. Capacitors C1...C4, C8 - K73-17, C5, C6, C9, SY - K50-24, the rest - KM-6. The KS212K zener diode can be replaced with KS212Zh or KS512A. Diodes VD8, VD9 must be installed on radiators with a dissipation area of at least 20 cm2 each. The efficiency of both blocks can be increased if, instead of KD213A diodes, Schottky diodes are used, for example, any of the KD2997 series. In this case, heat sinks for diodes will not be required.

Single-ended converters with high efficiency, 12/220 volts

Some common household electrical appliances, such as a lamp daylight, photo flash and a number of others, sometimes it is convenient to use in a car.

Since most devices are designed to be powered from a network with an operating voltage of 220 V, a step-up converter is needed. An electric razor or a small fluorescent lamp consumes no more than 6...25 W of power. Moreover, such a converter is often not required AC voltage at the exit. The above household electrical appliances operate normally when powered by direct or unipolar pulsating current.

The first version of a single-cycle (flyback) pulsed DC voltage converter 12 V/220 V is made on an imported UC3845N PWM controller chip and a powerful N-channel field-effect transistor BUZ11 (Fig. 4.10). These items are more affordable than domestic analogues, and make it possible to achieve high efficiency from the device, including due to the low source-drain voltage drop across the open field-effect transistor (the efficiency of the converter also depends on the ratio of the width of the pulses transmitting energy to the transformer to the pause).

The specified microcircuit is specially designed for single-cycle converters and has all the necessary components inside, which allows reducing the number of external elements. It has a high-current quasi-complementary output stage specifically designed for direct power control. M-channel field-effect transistor with insulated gate. The operating pulse frequency at the output of the microcircuit can reach 500 kHz. The frequency is determined by the ratings of elements R4-C4 and in the above circuit is about 33 kHz (T = 50 μs).

Rice. 4.10. Circuit of a single-cycle pulse converter that increases voltage

The chip also contains a protection circuit to shut down the converter when the supply voltage drops below 7.6 V, which is useful when powering devices from a battery.

Let's take a closer look at the operation of the converter. In Fig. Figure 4.11 shows voltage diagrams that explain the ongoing processes. When positive pulses appear on the gate field effect transistor(Fig. 4.11, a) it opens and on resistors R7-R8 there will be pulses shown in Fig. 4.11, c.

The slope of the top of the pulse depends on the inductance of the transformer winding, and if at the top there is a sharp increase in the voltage amplitude, as shown by the dotted line, this indicates saturation of the magnetic circuit. At the same time, conversion losses increase sharply, which leads to heating of the elements and deteriorates the operation of the device. To eliminate saturation, you will need to reduce the pulse width or increase the gap in the center of the magnetic circuit. Usually a gap of 0.1...0.5 mm is sufficient.

When the power transistor is turned off, the inductance of the transformer windings causes voltage surges to appear, as shown in the figures.

Rice. 4.11. Voltage diagrams at circuit control points

With the correct manufacture of transformer T1 (sectioning the secondary winding) and low-voltage power supply, the surge amplitude does not reach a value dangerous for the transistor and therefore in this circuit special measures, in the form of damping circuits in the primary winding T1, is not used. And in order to suppress surges in the current feedback signal coming to the input of the DA1.3 microcircuit, a simple RC filter from elements R6-C5 is installed.

The voltage at the converter input, depending on the condition of the battery, can vary from 9 to 15 V (which is 40%). To limit the change in output voltage, input feedback is removed from the divider of resistors R1-R2. In this case, the output voltage at the load will be maintained in the range of 210...230 V (Rload = 2200 Ohm), see table. 4.2, i.e. it changes by no more than 10%, which is quite acceptable.

Table 4.2. Circuit parameters when changing supply voltage

Stabilization of the output voltage is carried out by automatically changing the width of the pulse that opens transistor VT1 from 20 μs at Upit = 9 V to 15 μs (Upit = 15 V).

All elements of the circuit, except for capacitor C6, are placed on a single-sided printed circuit board made of fiberglass with dimensions of 90x55 mm (Fig. 4.12).

Rice. 4.12. Topology printed circuit board and arrangement of elements

Transformer T1 is mounted on the board using an M4x30 screw through a rubber gasket, as shown in Fig. 4.13.

Rice. 4.13 Mounting type of transformer T1

Transistor VT1 is installed on the radiator. Plug design. XP1 must prevent erroneous supply of voltage to the circuit.

The T1 pulse transformer is made using the widely used BZO armor cups from the M2000NM1 magnetic core. At the same time, in the central part they should have a gap of 0.1...0.5 mm.

The magnetic core can be purchased with an existing gap or it can be made using coarse sandpaper. It is better to select the gap size experimentally when tuning so that the magnetic circuit does not enter the saturation mode - this is convenient to control by the shape of the voltage at the source VT1 (see Fig. 4.11, c).

For transformer T1, winding 1-2 contains 9 turns of wire with a diameter of 0.5-0.6 mm, windings 3-4 and 5-6 each contain 180 turns of wire with a diameter of 0.15...0.23 mm (wire type PEL or PEV). In this case, the primary winding (1-2) is located between two secondary windings, i.e. First, winding 3-4 is wound, and then 1-2 and 5-6.

When connecting the transformer windings, it is important to observe the phasing shown in the diagram. Incorrect phasing will not damage the circuit, but it will not work as intended.

The following parts were used during assembly: adjusted resistor R2 - SPZ-19a, fixed resistors R7 and R8 type S5-16M for 1 W, the rest can be of any type; electrolytic capacitors C1 - K50-35 for 25 V, C2 - K53-1A for 16 V, C6 - K50-29V for 450 V, and the rest are of the K10-17 type. Transistor VT1 is installed on a small (by the size of the board) radiator made of duralumin profile. Setting up the circuit consists of checking the correct phrasing of connecting the secondary winding using an oscilloscope, as well as setting resistor R4 to the desired frequency. Resistor R2 sets the output voltage at the XS1 sockets when the load is on.

The given converter circuit is designed to work with a previously known load power (6...30 W - permanently connected). At idle, the voltage at the circuit output can reach 400 V, which is not acceptable for all devices, as it can lead to damage due to insulation breakdown.

If the converter is intended to be used in operation with a load of different power, which is also turned on during operation of the converter, then it is necessary to remove the voltage feedback signal from the output. A variant of such a scheme is shown in Fig. 4.14. This not only allows you to limit the output voltage of the circuit in idle mode to 245 V, but also reduces the power consumption in this mode by about 10 times (Ipot=0.19 A; P=2.28 W; Uh=245 V).

Rice. 4.14. Single-cycle converter circuit with maximum no-load voltage limitation

Transformer T1 has the same magnetic circuit and winding data as in the circuit (Fig. 4.10), but contains an additional winding (7-4) - 14 turns of PELSHO wire with a diameter of 0.12.0.18 mm (it is wound last). The remaining windings are made in the same way as in the transformer described above.

To manufacture a pulse transformer, you can also use square cores of the series. KV12 made of M2500NM ferrite - the number of turns in the windings in this case will not change. To replace armor magnetic cores (B) with more modern square ones (KB), you can use the table. 4.3.

The voltage feedback signal from winding 7-8 is supplied through a diode to the input (2) of the microcircuit, which makes it possible to more accurately maintain the output voltage in a given range, as well as provide galvanic isolation between the primary and output circuits. The parameters of such a converter, depending on the supply voltage, are given in table. 4.4.

Table 4.4. Circuit parameters when changing supply voltage

The efficiency of the described converters can be increased a little more if the pulse transformers are secured to the board with a dielectric screw or heat-resistant glue. A variant of the printed circuit board topology for assembling the circuit is shown in Fig. 4.15.

Rice. 4.15. PCB topology and arrangement of elements

Using such a converter, you can power electric shavers "Agidel", "Kharkov" and a number of other devices from the vehicle's on-board network.

Today we will look at several circuits of simple, one might even say simple, pulse DC-DC voltage converters (DC voltage converters of the same magnitude, in constant pressure different size)

What are the benefits of pulse converters? Firstly, they have high efficiency, and secondly, they can operate at an input voltage lower than the output voltage. Pulse converters are divided into groups:

- bucking, boosting, inverting;
- stabilized, unstabilized;
- galvanically isolated, non-insulated;
- with a narrow and wide range of input voltages.

To make homemade pulse converters, it is best to use specialized integrated circuits - they are easier to assemble and not capricious when setting up. So, here are 14 schemes for every taste:

This converter operates at a frequency of 50 kHz, galvanic isolation is provided by transformer T1, which is wound on a K10x6x4.5 ring made of 2000NM ferrite and contains: primary winding - 2x10 turns, secondary winding - 2x70 turns of PEV-0.2 wire. Transistors can be replaced with KT501B. Almost no current is consumed from the battery when there is no load.

Transformer T1 is wound on a ferrite ring with a diameter of 7 mm, and contains two windings of 25 turns of wire PEV = 0.3.

Push-pull unstabilized converter based on a multivibrator (VT1 and VT2) and a power amplifier (VT3 and VT4). The output voltage is selected by the number of turns of the secondary winding of the pulse transformer T1.

Stabilizing type converter based on the MAX631 microcircuit from MAXIM. Generation frequency 40…50 kHz, storage element - inductor L1.

You can use one of the two chips separately, for example the second one, to multiply the voltage from two batteries.

Typical circuit for connecting a pulse boost stabilizer on the MAX1674 microcircuit from MAXIM. Operation is maintained at an input voltage of 1.1 volts. Efficiency - 94%, load current - up to 200 mA.

Allows you to obtain two different stabilized voltages with an efficiency of 50...60% and a load current of up to 150 mA in each channel. Capacitors C2 and C3 are energy storage devices.

8. Pulse boost stabilizer on the MAX1724EZK33 chip from MAXIM

Typical circuit diagram for connecting a specialized microcircuit from MAXIM. It remains operational at an input voltage of 0.91 volts, has a small-sized SMD housing and provides a load current of up to 150 mA with an efficiency of 90%.

A typical circuit for connecting a pulsed step-down stabilizer on a widely available TEXAS microcircuit. Resistor R3 regulates the output voltage within +2.8…+5 volts. Resistor R1 sets the current short circuit, which is calculated by the formula: Ikz(A)= 0.5/R1(Ohm)